Passive switched-capacitor filters conforming to power constraint

ABSTRACT

Passive switched-capacitor (PSC) filters are described herein. In one design, a PSC filter implements a second-order infinite impulse response (IIR) filter with two complex first-order IIR sections. Each complex first-order IIR section includes three sets of capacitors. A first set of capacitors receives a real input signal and an imaginary delayed signal, stores and shares electrical charges, and provides a real filtered signal. A second set of capacitors receives an imaginary input signal and a real delayed signal, stores and shares electrical charges, and provides an imaginary filtered signal. A third set of capacitors receives the real and imaginary filtered signals, stores and shares electrical charges, and provides the real and imaginary delayed signals. In another design, a PSC filter implements a finite impulse response (FIR) section and an IIR section for a complex first-order IIR section. The IIR section includes multiple complex filter sections operating in an interleaved manner.

CROSS-REFERENCE TO RELATED APPLICATION

The present application for patent is a divisional of patent application Ser. No. 12/366,363 entitled “PASSIVE SWITCHED-CAPACITOR FILTERS CONFORMING TO POWER CONSTRAINT” filed Feb. 5, 2009, pending, and assigned to the assignee hereof and hereby expressly incorporated by reference herein.

BACKGROUND

I. Field

The present disclosure relates generally to electronics, and more specifically to filters.

II. Background

Filters are commonly used to filter signals to pass desired signal components and to attenuate undesired signal components. Filters are widely used for various applications such as communication, computing, networking, consumer electronics, etc. For example, in a wireless communication device such as a cellular phone, filters may be used to filter a received signal to pass a desired signal on a specific frequency channel and to attenuate out-of-band undesired signals and noise. For many applications, filters that occupy small area and consume low power are highly desirable.

SUMMARY

Passive switched-capacitor (PSC) filters that may occupy smaller area and consume less power are described herein. In one design, a PSC filter may implement a second-order infinite impulse response (IIR) filter with two complex first-order IIR sections. The second-order IIR filter may not meet a power constraint whereas each complex first-order IIR section may meet the power constraint. The coefficients for the two complex first-order IIR sections may be determined based on the coefficients for the second-order IIR filter, as described below. Each complex first-order IIR section may be implemented with a PSC filter section comprising multiple capacitors and multiple switches.

In another design, a PSC filter may implement one or more complex filter sections (e.g., two complex first-order IIR sections) coupled in series. Each complex filter section includes first, second, and third sets of capacitors. The first set of capacitors (e.g., capacitors 1024 a and 1034 a in FIG. 10) receives a real input signal and an imaginary delayed signal, stores and shares electrical charges, and provides a real filtered signal. The second set of capacitors (e.g., capacitors 1024 b and 1034 b in FIG. 10) receives an imaginary input signal and a real delayed signal, stores and shares electrical charges, and provides an imaginary filtered signal. The third set of capacitors (e.g., capacitors 1044 and 1054 in FIG. 10) receives the real and imaginary filtered signals, stores and shares electrical charges, and provides the real and imaginary delayed signals. Each complex filter section further includes first, second, third and fourth sets of switches. The first set of switches couples the first set of capacitors to a first summing node. The second set of switches couples the second set of capacitors to a second summing node. The third set of switches couples the third set of capacitors to the first summing node. The fourth set of switches couples the third set of capacitors to the second summing node. Each capacitor stores a value from an associated summing node when selected for charging and shares electrical charge with other capacitors via the associated summing node when selected for charge sharing.

In yet another design, a PSC filter may implement a finite impulse response (FIR) section coupled to an IIR section, which may be for a complex first-order IIR filter. The FIR section receives and filters a complex input signal and provides a complex filtered signal. The IIR section receives and filters the complex filtered signal and provides a complex output signal. The FIR and IIR sections may be implemented with two PSC filter sections. Each PSC filter section may include a bank of complex filter sections that may be enabled in different clock cycles.

In yet another design, a PSC filter section includes first and second complex filter sections and may be used for the FIR or IIR section described above. The first complex filter section receives and filters a complex input signal and provide a complex output signal every M clock cycles, where M is greater than one. The second complex filter section receives and filters the complex input signal and provides the complex output signal every M clock cycles. The first and second complex filter sections may be enabled in different clock cycles. For example, with M=2, the first complex filter section may be enabled in even-numbered clock cycles, and the second complex filter section may be enabled in odd-numbered clock cycles.

Various aspects and features of the disclosure are described in further detail below.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 shows a block diagram of a second-order FIR filter.

FIG. 2 shows a PSC filter that implements the second-order FIR filter.

FIG. 3 shows a timing diagram for the PSC filter in FIG. 2.

FIG. 4 shows a block diagram of a second-order IIR filter.

FIG. 5 shows a PSC filter that implements the second-order IIR filter.

FIG. 6 shows a timing diagram for the PSC filter in FIG. 5.

FIG. 7 shows a process for designing a PSC filter with coefficient scaling.

FIG. 8 shows filtering with complex second-order IIR filters.

FIG. 9 shows a block diagram of a complex first-order IIR section.

FIG. 10 shows a PSC filter that implements the complex first-order IIR section.

FIG. 11 shows a timing diagram for the PSC filter in FIG. 10.

FIG. 12 shows a PSC filter that implements two complex first-order IIR sections.

FIG. 13 shows a process for designing a PSC filter with decomposition.

FIG. 14 shows a complex first-order IIR section implemented with a FIR filter bank and an IIR filter bank.

FIG. 15 shows a PSC filter that implements the IIR filter bank in FIG. 14.

FIG. 16 shows a plot of pole movement due to filter bank transformation.

FIG. 17 shows a process for designing a PSC filter with filter bank transformation.

FIG. 18 shows a plot of a function affecting power constraint.

FIG. 19 shows a process for designing an IIR filter to meet power constraint.

FIG. 20 shows a block diagram of a wireless communication device.

DETAILED DESCRIPTION

The PSC filters described herein may be used for various types of filters such as FIR filters, IIR filters, auto regressive moving average (ARMA) filters composed of FIR and IIR sections, etc. The PSC filters may also implement a filter of any order, e.g., first, second, third or higher order. Multiple PSC filter sections may be used to form more complex filters. For clarity, PSC filters for second-order FIR filter and for first-order and second-order IIR filters are described in detail below.

A PSC filter may be implemented with only capacitors and switches, without using active circuits. This may provide certain advantages described below. However, due to the passive nature of the PSC filter, not all filter transfer functions may be directly implementable with the PSC filter. The PSC filter can implement a filter transfer function that meets a power constraint. Various schemes to meet the power constraint are described below and are based on observation that the total electrical charges before and after each charge sharing operation in the PSC circuit should be keep constant. This implies that a FIR filter is implementable if its coefficients are scaled so that they sum to 1. For an IIR filter, several schemes for meeting the power constraint are described below and include coefficient scaling, complex filter section decomposition, filter bank transformation, and pole repositioning.

FIG. 1 shows a block diagram of a second-order FIR filter 100 that may be implemented with a PSC filter. FIR filter 100 includes two delay elements 112 and 114 coupled in series, with each delay element providing a delay of one clock cycle. Delay element 112 receives an input sample x(n) and provides a delayed sample x(n−1). Delay element 114 receives the delayed sample x(n−1) and provides a delayed sample x(n−2). FIR filter 100 includes two FIR taps 1 and 2 for second order. A multiplier 120 (which may be considered as being for FIR tap 0) is coupled to the input of delay element 112. A multiplier 122 for FIR tap 1 is coupled to the output of delay element 112. A multiplier 124 for FIR tap 2 is coupled to the output of delay element 114. Multipliers 120, 122 and 124 multiply their samples with filter coefficients b₀, b₁ and b₂, respectively. A summer 130 is coupled to the outputs of all three multipliers 120, 122 and 124. Summer 130 sums the outputs of multipliers 120, 122 and 124 and provides an output sample y(n).

The output sample y(n) from FIR filter 100 may be expressed as:

y(n)=b ₀ ·x(n)+b ₁ ·x(n−1)+b ₂ ·x(n−2).  Eq (1)

A transfer function H_(FIR)(z) for FIR filter 100 in the z-domain may be expressed as:

H _(FIR)(z)=b ₀ +b ₁ ·z ⁻¹ +b ₂ ·z ⁻²,  Eq (2)

where z^(−k) denotes a delay of k clock cycles.

The filter coefficients may be defined to meet the following power constraint for FIR filter:

|b ₀ |+|b ₁ |+|b ₂|=1.  Eq (3)

If coefficients b₀, b₁ and b₂ do not meet the power constraint in equation (3), then the coefficients may be scaled as follows:

b ₀ ′=K _(FIR) ·b ₀,  Eq (4a)

b ₁ ′=K _(FIR) ·b ₁,and  Eq (4b)

b ₂ ′=K _(FIR) ·b ₂,  Eq (4c)

where

$K_{FIR} = \frac{1}{{b_{0}} + {b_{1}} + {b_{2}}}$

is a scaling factor.

Eq (4d)

The scaling in equation set (4) results in the scaled coefficients meeting the power constraint, as follows:

|b ₀ ′|+|b ₁ ′|+|b ₂′|=1.  Eq (5)

Any set of FIR filter coefficients may be scaled to meet the power constraint in equation (5). The power constraint may also be referred to as an absolute magnitude sum constraint. The FIR filter may be implemented with the scaled coefficients and may generate scaled output sample y′(n), which may be expressed as:

y′(n)=b ₀ ′·x(n)+b ₁ ′·x(n−1)+b ₂ ′·x(n−2)=K _(FIR) ·y(n).  Eq (6)

In many cases, y′(n) may be used in place of y(n). However, in cases where signal level plays a non-trivial role (e.g., to avoid signal saturation), the K_(FIR) scaling factor may be increased or decreased. Active devices such as amplifiers may be used to increase K_(FIR). A switching pattern may be adjusted to reduce K_(FIR).

FIG. 2 shows a schematic diagram of a design of a PSC filter 200 that implements second-order FIR filter 100 in FIG. 1. PSC filter 200 includes an input section 220 and two tap sections 230 and 240 for FIR taps 1 and 2, respectively, of FIR filter 100. Within PSC filter 200, an input switch 212 has one end receiving an input signal V_(in) and the other end coupled to a summing node A. A reset switch 214 is coupled between the summing node and circuit ground. An output switch 216 has one end coupled to the summing node and the other end providing an output signal V_(out). Switches 212, 214 and 216 may be implemented with metal oxide semiconductor (MOS) transistors or other types of transistors or switches.

Input section 220 includes an input capacitor 224 coupled between the summing node and circuit ground. Tap section 230 includes two switches 232 a and 232 b coupled in series with two capacitors 234 a and 234 b, respectively. Both series combinations of switch 232 and capacitor 234 are coupled between the summing node and circuit ground. Tap section 240 includes three switches 242 a, 242 b and 242 c coupled in series with three capacitors 244 a, 244 b and 244 c, respectively. All three series combinations of switch 242 and capacitor 244 are coupled between the summing node and circuit ground.

All capacitor(s) in each section have the same capacitance/size, which is determined by the corresponding filter coefficient. The capacitances of the capacitors in the three sections of PSC filter 200 may be given as:

C ₀₀ =K·b ₀′,  Eq (7a)

C ₁₀ =C ₁₁ =K·b ₁′,and  Eq (7b)

C ₂₀ =C ₂₁ =C ₂₂ =K·b ₂′,  Eq (7c)

where C_(ij) is the capacitance of the j-th capacitor in the section for FIR tap i, and

K is a scaling constant.

As shown in equation set (7), the size of each capacitor C_(ij) is proportional to the corresponding scaled coefficient b_(i)′. K may be selected based on various factors such as switching settling time, capacitor size, power dissipation, noise, etc. A negative capacitor for a negative coefficient may be obtained by switching the polarity of the capacitor between a read phase and a charge sharing phase.

In each clock cycle, switch 212 is closed for a brief period of time to charge one capacitor in each section with the V_(in) signal. The capacitor selected for charging in each tap section is determined by switches 232 and 242, as described below. The total input capacitance observed by the V_(in) signal for the charge operation may be expressed as:

C _(in) =C ₀₀ +C _(1u) +C _(2v),  Eq (8)

where uε{0, 1} is an index of the capacitor selected for charging in tap section 230, and

vε{0, 1, 2} is an index of the capacitor selected for charging in tap section 240.

Since the capacitors in each tap section have the same capacitance, the total input capacitance C_(in) is constant for each clock cycle.

In each clock cycle, an appropriate capacitor in each tap section is used to generate the V_(out) signal. For FIR tap L, the capacitor charged L clock cycles earlier and storing x(n−L) is selected for use via its associated switch. The two selected capacitors in tap sections 230 and 240 and input capacitor 224 are used in a charge sharing operation that implements the multiplications with filter coefficients b₀′ through b₂′ and the summing of the multiplier outputs in equation (6).

The charge sharing operation uses capacitor size to achieve multiplication with a filter coefficient and current summing to achieve summing of the multiplier outputs. For each capacitor within PSC filter 200, the voltage V_(ij) across that capacitor is determined by the V_(in) signal at the time the capacitor is charged, or V_(ij)=V_(in). The electrical charge Q_(ij) stored by each capacitor is determined by the voltage V_(ij) across that capacitor and the capacitance C_(ij) of the capacitor, or Q_(ij)=V_(ij)·C_(ij). In each clock cycle, one capacitor storing the proper sample x(n−1) from each tap section is selected, and the charges from all selected capacitors as well as input capacitor 224 are shared. The charge sharing for the FIR filter may be expressed as:

$\begin{matrix} {{V_{out} = \frac{{C_{00} \cdot V_{00}} + {C_{1p} \cdot V_{1p}} + {C_{2q} \cdot V_{2q}}}{C_{00} + C_{1p} + C_{2q}}},} & {{Eq}\mspace{14mu} (9)} \end{matrix}$

-   where pε{0, 1} is an index of the capacitor storing x(n−1) in tap     section 230, and qε{0, 1, 2} is an index of the capacitor storing     x(n−2) in tap section 240.

Since the capacitors in each tap section have the same capacitance, the total output capacitance C_(out) observed by the V_(out) signal is constant for each clock cycle and is equal to the total input capacitance, or C_(out)=C_(in).

Index p can cycle between 0 and 1, so that in each clock cycle one capacitor 234 in tap section 230 is charged, and the other capacitor 234 is used for charge sharing. Index q can cycle from 0 through 2, so that in each clock cycle one capacitor 244 in tap section 240 is charged, and another capacitor 244 is used for charge sharing. PSC filter 200 may be considered as having six states for the six different (p, q) combinations.

FIG. 3 shows a timing diagram of various control signals for PSC filter 200 in FIG. 2. A clock signal CLK is shown at the top of the timing diagram. Control signals for the switches within PSC filter 200 are shown below the clock signal.

In the design shown in FIG. 3, each clock cycle includes a read/charge phase, a compute/charge sharing phase, a write/output phase, and a reset/discharge phase. For the read phase from time T₀ to time T₁, the S_(in) control signal is asserted, switch 212 is closed, and input capacitor C₀₀ and one capacitor in each tap section are charged with the V_(in) signal. The S_(ij) control signal for each capacitor selected for charging is asserted during the read phase and de-asserted at time T₂. For the charge sharing phase starting at time T₃, the S_(ij) control signal for each capacitor selected for charge sharing is asserted, and the selected capacitors in sections 230 and 240 as well as capacitor C₀₀ perform charge sharing via the summing node. For the write phase from time T₄ to time T₅, the S_(out) control signal is asserted, switch 216 is closed, and the voltage at the summing node is provided as the K_(out) signal. For the reset phase from time T₆ to time T₇, the S_(reset) control signal is asserted, switch 214 is closed, and the capacitors used for charge sharing are reset/discharged. These capacitors may be charged with the V_(in) signal in the next clock cycle.

FIG. 4 shows a block diagram of a second-order IIR filter 400 that may be implemented with a PSC filter. Within IIR filter 400, a multiplier 420 receives and scales an input sample x(n) with a filter coefficient c₀. A summer 430 subtracts the output of a summer 432 from the output of multiplier 420 and provides an output sample y(n).

Two delay elements 412 and 414 are coupled in series, with each delay element providing a delay of one clock cycle. Delay element 412 receives the output sample y(n) and provides a delayed sample y(n−1). Delay element 414 receives delayed sample y(n−1) and provides a delayed sample y(n−2). IIR filter 400 includes two IIR taps 1 and 2 for second order. A multiplier 422 for IIR tap 1 is coupled to the output of delay element 412. A multiplier 424 for IIR tap 2 is coupled to the output of delay element 414. Multipliers 422 and 424 multiply their samples with filter coefficients c₁ and c₂, respectively, for the two IIR taps. Summer 432 sums the outputs of multipliers 422 and 424 and provides its output to summer 430.

The output sample y(n) from IIR filter 400 may be expressed as:

y(n)=c ₀ ·x(n)−c ₁ ·y(n−1)−c ₂ ·y(n−2).  Eq (10)

A transfer function H_(IIR)(z) for IIR filter 400 may be expressed as:

$\begin{matrix} {{H_{IIR}(z)} = {\frac{c_{0}}{1 + {c_{1} \cdot z^{- 1}} + {c_{2} \cdot z^{- 2}}}.}} & {{Eq}\mspace{14mu} (11)} \end{matrix}$

FIG. 5 shows a schematic diagram of a design of a PSC filter 500 that implements second-order IIR filter 400 in FIG. 4. PSC filter 500 includes an input section 520 and two tap sections 530 and 540 for IIR taps 1 and 2, respectively, of IIR filter 400. Within PSC filter 500, an input switch 512 has one end receiving an input signal V_(in) and the other end coupled to a summing node A. A reset switch 514 is coupled between the summing node and circuit ground. An output switch 516 has one end coupled to the summing node and the other end providing an output signal V_(out).

Input section 520 includes a capacitor 524 coupled between the summing node and circuit ground. Tap section 530 includes a switch 532 coupled in series with a capacitor 534, the combination of which is coupled between the summing node and circuit ground. Tap section 540 includes two switches 542 a and 542 b coupled in series with two capacitors 544 a and 544 b, respectively. Both series combinations of switch 542 and capacitor 544 are coupled between the summing node and circuit ground. Capacitors 534 and 544 in tap sections 530 and 540 may be reset at the start of filtering operation.

All capacitor(s) in each section of PSC filter 500 have the same capacitance, which is determined by the corresponding filter coefficient. The capacitances of the capacitors in the three sections of PSC filter 500 may be given as:

C ₀₀ =K·c ₀,  Eq (12a)

C ₁₀ =K·c ₁,and  Eq (12b)

C ₂₀ =C ₂₁ =K·c ₂.  Eq (12c)

As shown in equation set (12), the size of each capacitor C_(ij) is proportional to the magnitude of the corresponding filter coefficient c₁. A negative capacitor for a negative coefficient may be obtained by switching the polarity of the capacitor between the read phase and the charge sharing phase.

In each clock cycle, switch 512 is closed for a brief period of time to charge capacitor 524 in section 520 with the V_(in) signal. The total input capacitance observed by the V_(in) signal is thus C_(in)=C₀₀, and no extra capacitors are needed for C_(in).

In each clock cycle, an appropriate capacitor in each tap section is used to generate the V_(out) signal. For IIR tap L, the capacitor charged L clock cycles earlier and storing y(n−L) is selected for use via its associated switch. Two selected capacitors in tap sections 530 and 540 as well as input capacitor 524 are used in a charge sharing operation that implements the multiplications with filter coefficients c₀ through c₂ and the summing of the multiplier outputs in equation (10). The charge sharing for the IIR filter may be expressed as:

$\begin{matrix} {{V_{out} = \frac{{C_{00} \cdot V_{00}} + {C_{10} \cdot V_{10}} + {C_{2k} \cdot V_{2k}}}{C_{00} + C_{10} + C_{2k}}},} & {{Eq}\mspace{14mu} (13)} \end{matrix}$

where kεE{0, 1} is an index of the capacitor storing y(n−2) in tap section 540.

After completing the charge sharing, the voltage across capacitors C₀₀, C₁₀ and C_(2k) corresponds to y(n). Capacitors C₁₀ and C_(2k) may store y(n) for use in subsequent clock cycles. Capacitor C₀₀ may provide y(n) for the V_(out) signal. The total output capacitance observed by the V_(out) signal is C_(out)=C₀₀, and no extra capacitors are needed for C_(out).

Index k can cycle between 0 and 1, so that each capacitor 544 in tap section 540 is used for charge sharing in alternating clock cycle. PSC filter 500 may be considered as having two states for the two possible values of k.

FIG. 6 shows a timing diagram of various control signals for PSC filter 500 in FIG. 5. The clock signal CLK is shown at the top of the timing diagram. Control signals for the switches within PSC filter 500 are shown below the clock signal.

In the design shown in FIG. 6, each clock cycle includes a read phase, a charge sharing phase, a write phase, and a reset phase. For the read phase from time T₀ to time T₁, the S_(in) control signal is asserted, switch 512 is closed, and input capacitor C₀₀ is charged with the V_(in) signal. For the charge sharing phase from time T₂ to time T₃, the S_(ij) control signal for each capacitor selected for charge sharing is asserted, and the selected capacitors as well as input capacitor C₀₀ perform charge sharing via the summing node. At the end of the charge sharing phase, the S_(ij) control signal for each selected capacitor is de-asserted at time T₃, which then causes that capacitor to store y(n). For the write phase from time T₄ to time T₅, the S_(out) control signal is asserted, switch 516 is closed, and capacitor C₀₀ provides y(n) to the V_(out) signal. For the reset phase from time T₆ to time T₇, the S_(reset) control signal is asserted, switch 514 is closed, and capacitor C₀₀ is reset.

The coefficients for the second-order IIR filter may be defined to meet the following power constraint for IIR filter:

|c ₀ |+|c ₁ |c ₂|=1.  Eq (14)

If coefficients c₀′, c₁ and c₂ do not meet the power constraint in equation (14), then several schemes may be used meet the power constraint.

In a first scheme for meeting the power constraint for IIR filter, if |c₁|+|c₂|<1, then a scaled coefficient c₀′ may be computed as follows:

c ₀′=1−|c ₁ |−|c ₂|.  Eq (15)

The coefficients c₀′, c₁ and c₂ meet the power constraint for IIR filter, as follows:

|c ₀ ′|+|c ₁ |+|c ₂|=1.  Eq (16)

The IIR filter may be implemented with coefficients c₀′, c₁ and c₂ and may generate output sample y′(n), which may be expressed as:

y′(n)=c ₀ ′·x(n)−c ₁ ·y(n−1)−c ₂ ·y(n−2).  Eq (17)

The transfer function with coefficients c₀′, c₁ and c₂ may be expressed as:

$\begin{matrix} {{H_{IIR}^{\prime}(z)} = {{\frac{c_{0}^{\prime}}{c_{0}} \cdot {H_{IIR}(z)}} = {\frac{1 - {c_{1}} - {c_{2}}}{c_{0}} \cdot {{H_{IIR}(z)}.}}}} & {{Eq}\mspace{14mu} (18)} \end{matrix}$

FIG. 7 shows a process 700 for designing a PSC filter with coefficient scaling to meet the power constraint. Multiple coefficients for a filter transfer function may be obtained (block 712). At least one of the multiple coefficients may be scaled based on a power constraint for the PSC filter (block 714). The PSC filter may then be implemented based on the at least one scaled coefficient to obtain the filter transfer function (block 716).

In one design, the filter transfer function may be for a FIR filter, e.g., a second-order FIR filter having the power constraint shown in equation (3). In this design, a scaling factor K_(m) may be determined based on the magnitude of each of the multiple coefficients, e.g., as shown in equation (4d). Each of the multiple coefficients may then be scaled based on the scaling factor to obtain a corresponding scaled coefficient, e.g., as shown in equations (4a) through (4c).

In another design, the filter transfer function may be for an IIR filter, e.g., a second-order IIR filter having the power constraint shown in equation (14). In this design, one of the multiple coefficients may be replaced with a new coefficient determined based on magnitude of each remaining coefficient, e.g., as shown in equation (15).

In a second scheme for meeting the power constraint for IIR filter, if |c₁|+|c₂|≧1, then the second-order IIR filter may be decomposed into two first-order IIR sections. Lower-order IIR sections often (but not always) result in smaller coefficients, which may allow the power constraint to be met.

The decomposition of a second-order FIR section may be expressed as:

$\quad\begin{matrix} \begin{matrix} {{1 + {c_{1} \cdot z^{- 1}} + {c_{2} \cdot z^{- 2}}} = {\left( {1 - {p \cdot z^{- 1}}} \right) \cdot \left( {1 - {p^{*} \cdot z^{- 1}}} \right)}} \\ {= {\left( {1 - {\left( {p_{re} + {j\; p_{im}}} \right) \cdot z^{- 1}}} \right) \cdot}} \\ {\left( {1 - {\left( {p_{re} - {j\; p_{im}}} \right) \cdot z^{- 1}}} \right)} \end{matrix} & {{Eq}\mspace{14mu} (19)} \end{matrix}$

where p=p_(re)+j p_(im) is a complex coefficient, c₁=−2p_(re), c₂=p_(re) ²+p_(im) ², and “*” denotes a complex conjugate.

As shown in equation (19), the decomposition of a second-order FIR section typically produces two complex first-order FIR sections with conjugated coefficients p and p*.

The complex coefficient may be tested for the following power constraint condition:

|p _(re) |+|p _(im)|<1.  Eq (20)

If the condition in equation (20) is satisfied, then the second-order IIR filter in equation (10) may be implemented with two concatenated complex first-order IIR sections, both of which meet the power constraint. A complex output sample y′(n) from the first complex first-order IIR section may be expressed as:

y _(re)′(n)={tilde over (c)} ₀ x _(re)(n)+p _(re) ·y _(re)′(n−1)−p _(im) ·y _(im)′(n−1),and  Eq (21a)

y _(im)′(n)={tilde over (c)} ₀ ·x _(im)(n)+p _(re) ·y _(im)′(n−1)+p _(im) ·y _(re)′(n−1),  Eq (21b)

where

x(n)=x_(re)(n)+j x_(im)(n) is a complex input sample,

y′(n)=y_(re)′(n)+j y_(im)′(n) is a complex output sample from the first section, and

{tilde over (c)}₀ is a scaled coefficient that may be given as:

{tilde over (c)} ₀=1−|p _(re) |−|p _(im)|.  Eq (22)

A complex output sample y″(n) from the second complex first-order IIR section may be expressed as:

y _(re)″(n)={tilde over (c)} ₀ ·y _(re)′(n)+p _(re) ·y _(re)″(n−1)+p _(im) ·y _(im)″(n−1),and  Eq (23a)

y _(im)″(n)={tilde over (c)} ₀ ·y _(im)′(n)+p _(re) ·y _(im)″(n−1)−p _(im) ·y _(re)″),  Eq (23b)

where y″(n)=y_(re)″(n)+j y_(im)″(n) is a complex output sample from the second section.

As shown in equation sets (21) and (23), the first and second complex first-order IIR sections have the same coefficients. The only difference in the two complex first-order IIR sections is the sign of the samples scaled by p_(im).

As an example, a second-order IIR filter may have coefficients c₀=1, c₁=−0.25 and c₂=0.75. Since |c₁|+|c₂|=1, the first scheme for scaling coefficients does not apply. Using the second scheme, second-order IIR filter may be decomposed into two complex first-order IIR sections with p_(re)=0.125, p_(im)=0.857, and {tilde over (c)}₀=0.018. Since |p_(re)|+|p_(im)|<1, the two complex first-order IIR sections meet the power constraint.

FIG. 8 shows filtering of complex input samples with complex second-order IIR filters 810 and 830. For IIR filter 810, real input samples x_(re)(n) may be filtered with a real second-order IIR filter 820 a, e.g., as shown in equation (10), to obtain real output samples y_(re)(n). Imaginary input samples x_(im)(n) may be filtered with a real second-order IIR filter 820 b to obtain imaginary output samples y_(im)(n). IIR filters 820 a and 820 b independently filter the real and imaginary parts of the complex input samples. IIR filters 820 a and 820 b are identical and have the same coefficients. However, IIR filters 820 a and 820 b may not meet the power constraint and thus may not be directly implementable.

For IIR filter 830, the complex input samples x_(re)(n) and x_(im)(n) may be filtered with a complex first-order IIR section 840 a, e.g., as shown in equation set (21), to obtain complex filtered samples y_(re)′(n) and y_(im)′(n). The complex filtered samples may be further filtered with a complex first-order IIR section 840 b, e.g., as shown in equation set (23), to obtain complex output samples y_(re)″(n) and y_(im)″(n).

IIR filter 810 composed of two real second-order IIR filters 820 a and 820 b for the real and imaginary parts is equivalent to IIR filter 830 composed of two complex first-order IIR sections 840 a and 840 b. The complex output samples y_(re)″(n) and y from IIR filter 830 are equivalent to the complex output samples y_(re)″(n) and y_(im)″(n) from IIR filter 810. However, complex first-order IIR sections 840 a and 840 b may be implementable whereas real second-order IIR filters 820 a and 820 b may not be implementable.

FIG. 9 shows a block diagram of complex first-order IIR section 840 a, which includes an IIR section 910 a for the real part and an IIR section 910 b for the imaginary part. Within IIR section 910 a, a multiplier 920 a receives and scales a real input sample x_(re)(n) with filter coefficient {tilde over (c)}₀. A summer 930 a sums the output of multiplier 920 a with the output of a summer 932 a and provides a real output sample y_(re)′. A delay element 912 a receives the real output sample y_(re)′ (n) and provides a real delayed sample y_(re)′(n−1). A multiplier 922 a for IIR tap A is coupled to the output of delay element 912 a. A multiplier 924 a for IIR tap B is also coupled to the output of delay element 912 a. Multipliers 922 a and 924 a multiply the real delayed sample y_(re)′(n−1) with filter coefficients p_(re) and p_(im), respectively, for IIR taps A and B. Summer 932 a subtracts the output of a multiplier 924 b in IIR section 910 b from the output of multiplier 922 a in IIR section 910 a and provides its output to summer 930 a.

IIR section 910 b includes all of the elements in IIR section 910 a. The elements in IIR section 910 b are coupled in the same way as the elements in IIR section 910 a with the following differences. A multiplier 922 b for IIR tap C and a multiplier 924 b for IIR tap D multiply the imaginary delayed sample y_(im)′(n−1) from a delay element 912 b with filter coefficients p_(re) and p_(im), respectively. A summer 932 b sums the output of multiplier 924 a in IIR section 910 a with the output of multiplier 922 b in IIR section 910 b and provides its output to a summer 930 b.

Complex first-order IIR section 840 b in FIG. 8 may be implemented in similar manner as complex first-order IIR section 840 a in FIG. 8, with the difference being a swamp in the signs of the outputs of multipliers 924 a and 924 b.

FIG. 10 shows a schematic diagram of a design of a PSC filter 1000 that implements complex first-order IIR section 840 a in FIGS. 8 and 9. PSC filter 1000 includes a path 1010 a for the real part and a path 1010 b for the imaginary part. Path 1010 a includes an input section 1020 a and a tap section 1030 a for IIR tap A of IIR section 840 a in FIG. 9. Path 1010 b includes an input section 1020 b and a tap section 1030 b for IIR tap C in FIG. 9. Both paths 1010 a and 1010 b share a tap section 1040 for IIR taps B and D in FIG. 9.

Within path 1010 a, an input switch 1012 a has one end receiving a real input signal V_(in,re) and the other end coupled to a summing node A. A reset switch 1014 a is coupled between summing node A and circuit ground. An output switch 1016 a has one end coupled to summing node A and the other end providing a real output signal V_(out,re). Switches 1012 b, 1014 b and 1016 b in path 1010 b are coupled in similar manner as switches 1012 a, 1014 a and 1016 a, respectively, in path 1010 a.

Input section 1020 a includes a capacitor 1024 a coupled between summing node A and circuit ground. Tap section 1030 a includes a switch 1032 a coupled in series with a capacitor 1034 a, the combination of which is coupled between summing node A and circuit ground. Input section 1020 b and tap section 1030 b are coupled in similar manner between summing node B and circuit ground. Tap section 1040 includes two switches 1042 a and 1052 a having one end coupled to summing node A and the other end coupled to capacitors 1044 and 1054, respectively. Tap section 1040 further includes two switches 1042 b and 1052 b having one end coupled to summing node B and the other end coupled to capacitors 1044 and 1054, respectively. The other ends of capacitors 1044 and 1054 are coupled to circuit ground.

The capacitances of the capacitors in PSC filter 1000 may be given as:

C ₀₀ =C ₀₁ =K· {tilde over (c)} ₀,  Eq (24a)

C ₁₀ =C ₁₁ =K· {tilde over (p)} _(re),and  Eq (24b)

C ₂₀ =C ₂₁ =K· p _(im).  Eq (24c)

For the example above with p_(re)=0.125, p_(im)=0.857, and {tilde over (c)}₀=0.018, the capacitance ratios may be given as follows:

C ₀₀ :C ₁₀ ;C ₁₀ :C ₁₁ ;C ₂₀ :C ₂₁ ={tilde over (c)} ₀ :{tilde over (c)} ₀ ;p _(re) :p _(re) ;p _(im) :p _(im)=0.018:0.018;0.125:0.125;0.857:0.857

In each clock cycle, switches 1012 a and 1012 b are closed for a brief period of time to charge capacitor 1024 a in section 1020 a with the V_(in,re) signal and to charge capacitor 1024 b in section 1020 b with the V_(in,im) signal. In each clock cycle, capacitor 1024 a in section 1020 a, capacitor 1034 a in section 1030 b, and either capacitor 1044 or 1054 in section 1040 are used in a charge sharing operation that implements the multiplications with filter coefficients {tilde over (c)}₀, p_(re) and p_(im), and the summing of the multiplier outputs in equation (21a). In each clock cycle, capacitor 1024 b in section 1020 b, capacitor 1034 b in section 1030 b, and either capacitor 1054 or 1044 in section 1040 are used in a charge sharing operation that implements equation (21b). The charge sharing for the real and imaginary parts may be expressed as:

$\begin{matrix} {{V_{{out},{re}} = \frac{{C_{00} \cdot V_{00}} + {C_{10} \cdot V_{10}} + {C_{2k} \cdot V_{2k}}}{C_{00} + C_{10} + C_{2k}}},{and}} & {{Eq}\mspace{14mu} \left( {25a} \right)} \\ {{V_{{out},{im}} = \frac{{C_{01} \cdot V_{01}} + {C_{11} \cdot V_{11}} + {C_{2\overset{\_}{k}} \cdot V_{2\overset{\_}{k}}}}{C_{01} + C_{11} + C_{2\overset{\_}{k}}}},} & {{Eq}\mspace{14mu} \left( {25b} \right)} \end{matrix}$

-   -   where kε{0, 1} is an index of the capacitor storing y_(im)′(n−1)         in tap section 1040, and k is an index of the capacitor storing         y_(re)′(n−1) in tap section 1040.

After completing the charge sharing, the voltage on summing node A corresponds to y_(re)′(n), and the voltage on summing node B corresponds to y_(im)′(n). Capacitor 1024 a may provide y_(re)′(n) for the V_(out,re) signal, and capacitor 1024 b may provide y_(im)′(n) for the V_(out,im) signal. Capacitors 1034 a may store y_(re)′(n) and capacitor 1034 b may store y_(im)′(n) for use in the next clock cycle. Capacitors 1044 and 1054 are used in an interleaved manner to store y_(re)′(n) and y_(im)′(n) in each clock cycle. In each even-numbered clock cycle, capacitor 1044 may be coupled to summing node A, perform charge sharing, and store the y_(re)′(n), while capacitor 1054 may be coupled to summing node B, perform charge sharing, and store the y_(im)′(n). In each odd-numbered clock cycle, capacitor 1044 may be coupled to summing node B, perform charge sharing, and store the y_(im)′(n), while capacitor 1054 may be coupled to summing node A, perform charge sharing, and store the y_(re)′(n). Capacitor 1044 may thus be coupled to summing nodes A and B in alternating clock cycles, and capacitor 1054 may be coupled to summing nodes B and A in alternating clock cycles. Capacitors 1044 and 1054 have the same size but are interleaved in time.

FIG. 6 shows a timing diagram that may be used for the various control signals for PSC filter 1000 in FIG. 10. For the read phase from time T₀ to time T₁, the S_(in) control signal is asserted, switches 1012 a and 1012 b are closed, capacitor C₀₀ is charged with the V_(in,re) signal, and capacitor C₀₁ is charged with the V_(in,im) signal. For the charge sharing phase from time T₂ to time T₃, the S₁₀ control signal and either the S₂₀ or S₂₂ control signal are asserted, and capacitor C₀₀, capacitor C₁₀ and either capacitor C₂₀ or C₂₂ perform charge sharing via summing node A. Simultaneously, the S₁₁ control signal and either the S₂₁ or S₂₃ control signal are asserted, and capacitor C₀₁, capacitor C₁₁ and either capacitor C₂₀ or C₂₂ perform charge sharing via summing node B. At the end of the charge sharing phase, the S_(ij) control signal for each selected capacitor is de-asserted at time T₃, which then causes that capacitor to store y_(re)′(n) or y_(im)′(n). For the write phase from time T₄ to time T₅, the S_(out) control signal is asserted, switches 1016 a and 1016 b are closed, capacitor C₀₀ provides y_(re)′(n) to the V_(out,re) signal, and capacitor C₀₁ provides y_(im)′(n) to the V_(out,im) signal. For the reset phase from time T₆ to time T₇, the S_(reset) control signal is asserted, switches 1014 a and 1014 b are closed, and capacitors C₀₀ and C₀₁ are reset.

FIG. 11 shows a timing diagram of a switching pattern for PSC filter 1000 in FIG. 10. The switching pattern includes two cycles 0 and 1 for the two possible values of k in equation set (25) and repeats every two clock cycles. Table 1 shows the two cycles 0 and 1 and, for each cycle, gives the capacitors used to generate the V_(out,re) and V_(out,im) signals.

TABLE 1 Cycle Capacitors used to Capacitors used to k generate V_(out, re) generate V_(out, im) 0 C₀₀, C₁₀ and C₂₀ C₀₁, C₁₁ and C₂₁ 1 C₀₀, C₁₀ and C₂₁ C₀₁, C₁₁ and C₂₀

For cycle 0, input capacitor C₀₀ is charged with the V_(in,re) signal and capacitor C₀₁ is charged with the V_(in,im) signal when the S_(in) control signal is asserted during the read phase. The S₁₀, S₁₁, S₂₀ and S₂₃ control signals are asserted during the charge sharing phase, capacitors C₀₀, C₁₀ and C₂₀ are used to generate the V_(out,re) signal, and capacitors C₀₁, C₁₁ and C₂₁ are used to generate the V_(out,im) signal. Capacitors C₁₀ and C₂₀ store the V_(out,re) signal and capacitors C₁₁ and C₂₁ store the V_(out,im) signal at the end of the charge sharing phase.

For cycle 1, input capacitor C₀₀ is charged with the V_(in,re) signal and capacitor C₀₁ is charged with the V_(in,im) signal during the read phase. The S₁₀, S₁₁, S₂₁ and S₂₂ control signals are asserted during the charge sharing phase, capacitors C₀₀, C₁₀ and C₂₁ are used to generate the V_(out,re) signal, and capacitors C₀₁, C₁₁ and C₂₀ are used to generate the V_(out,im) signal. Capacitors C₁₀ and C₂₁ store the V_(out,re) signal and capacitors C₁₁ and C₂₀ store the V_(out,im) signal at the end of the charge sharing phase.

For PSC filter 1000, capacitors C₀₀ and C₀₁ are charged with the V_(in,re) and V_(in,im) signals in each clock cycle and are also used for charge sharing in the same clock cycle. Capacitors C₁₀ and C₁₁ are used for charge sharing in each clock cycle and store y_(re)′(n) and y_(im)′(n) for use in the next clock cycle. For tap section 1040, index k toggles between 0 and 1, capacitors C₂₀ and C₂₁ are used for charge sharing at nodes A and B in one clock cycle, at nodes B and A in the following clock cycle, etc.

Table 2 summarizes the action performed by each capacitor in PSC filter 1000 in each clock cycle.

TABLE 2 Clock Cycle n n + 1 n + 2 . . . Capacitor C₀₀ store x_(re)(n) store x_(re)(n + 1) store x_(re)(n + 2) . . . x_(re)(n)→ y_(re)(n) x_(re)(n + 1)→ y_(re)(n + 1) x_(re)(n + 2)→ y_(re)(n + 2) Capacitor C₀₁ store x_(im)(n) store x_(im)(n + 1) store x_(im)(n + 2) . . . x_(im)(n)→ y_(im)(n) x_(im)(n + 1)→ y_(im)(n + 1) x_(im)(n + 2)→ y_(im)(n + 2) Capacitor C₁₀ y_(re)(n − 1)→ y_(re)(n) y_(re)(n)→ y_(re)(n + 1) y_(re)(n + 1)→ y_(re)(n + 2) . . . store y_(re)(n) store y_(re)(n + 1) store y_(re)(n + 2) Capacitor C₁₁ y_(im)(n − 1)→ y_(im)(n) y_(im)(n)→ y_(im)(n + 1) y_(im)(n + 1)→ y_(im)(n + 2) . . . store y_(im)(n) store y_(im)(n + 1) store y_(im)(n + 2) Capacitor C₂₀ y_(im)(n − 1)→ y_(re)(n) y_(re)(n)→ y_(im)(n + 1) y_(im)(n + 1)→ y_(re)(n + 2) . . . store y_(re)(n) store y_(im)(n + 1) store y_(re)(n + 2) Capacitor C₂₁ y_(re)(n − 1)→ y_(im)(n) y_(im)(n)→ y_(re)(n + 1) y_(re)(n + 1)→ y_(im)(n + 2) . . . store y_(im)(n) store y_(re)(n + 1) store y_(im)(n + 2)

FIG. 12 shows a schematic diagram of a design of a PSC filter 1200 that implements complex first-order IIR sections 840 a and 840 b in FIG. 8. PSC filter 1200 includes a first PSC filter section 1210 a that implements complex first-order IIR section 840 a and a second PSC filter section 1210 b that implements complex first-order IIR section 840 b. Each PSC filter section 1210 includes all elements in PSC filter 1000 in FIG. 10. The capacitors in tap section 1040 within PSC filter section 1210 a are selected in different order than the capacitors in tap section 1040 within PSC filter section 1210 b due to the difference between equation sets (21) and (23).

FIG. 13 shows a process 1300 for designing a PSC filter with decomposition. A filter transfer function may be decomposed into multiple complex first-order filter sections, e.g., as shown in equation (19) (block 1312). In one design of block 1312, the filter transfer function is for a second-order IIR filter and may be decomposed into two complex first-order IIR sections. Complex coefficients (e.g., p and p*) for the two complex first-order IIR sections may be determined based on coefficients (e.g., {tilde over (c)}₁ and c₂) for the second-order IIR filter. An input coefficient (e.g., c₀) for the two complex first-order IIR sections may be determined based on the magnitude of the real and imaginary parts (e.g., p_(re) and p_(im)) of the complex coefficients, e.g., as shown in equation (22). The multiple complex first-order filter sections may be implemented with multiple PSC filter sections to obtain the filter transfer function (block 1314).

The two complex first-order IIR sections obtained by decomposing a second-order IIR filter may not meet the power constraint. The complex pole obtained from the decomposition may be expressed as:

p=p _(re) +jp _(in) =r·e ^(jθ),  Eq (26)

where r is the magnitude of the pole and θ is the phase of the pole.

The sum of the magnitude of the real and imaginary parts of the pole may be expressed as:

≡p _(re) |+|p _(im) |=r·(|cos θ|+|sin θ|)  Eq (27)

A necessary and sufficient condition for stability of an IIR filter is r<1. The power constraint may be met with |p_(re)|+|p_(im)|<1. The term (|cos θ|+|sin θ|) may be greater than one depending on the value of θ. Thus, it is possible to have |p_(re)|+|p_(im)|≧1 even with r<1, in which case the IIR filter is stable but not directly implementable.

In a third scheme for meeting the power constraint for IIR filter, which may be used when the complex first-order IIR sections do not meet the power constraint, a complex first-order IIR section may be implemented with an interleaved filter bank. From equation set (21), the complex filtered samples from complex first-order IIR section 840 a may be expressed as:

$\quad\begin{matrix} \begin{matrix} {{y^{\prime}(n)} = {{{\overset{\sim}{c}}_{0} \cdot {x(n)}} + {p \cdot {y^{\prime}\left( {n - 1} \right)}}}} \\ {= {{{\overset{\sim}{c}}_{0} \cdot {x(n)}} + {p \cdot {\overset{\sim}{c}}_{0} \cdot {x\left( {n - 1} \right)}} + {p^{2} \cdot {{y^{\prime}\left( {n - 2} \right)}.}}}} \end{matrix} & {{Eq}\mspace{14mu} (28)} \end{matrix}$

Two consecutive filtered samples from complex first-order IIR section 840 a may be expressed as:

y′(n)={tilde over (c)} ₀ ·x(n)+p·{tilde over (c)} ₀ ·x(n−1)+p ² ·y′(n−2),and  Eq (29a)

y′(n+1)={tilde over (c)} ₀ ·x(n+1)+p·{tilde over (c)} ₀ ·x(n)+p ² ·y′(n−1).  Eq (29b)

Equation set (29) may be partitioned into an IIR part and a FIR part. The IIR part may also be referred to as a recursive part or an autoregressive part. The FIR part may also be referred to as a non-recursive part. The IIR part may be expressed as:

y′(n)={hacek over (c)} ₀ ·{hacek over (x)}(n)+p ² ·y′(n−2),and  Eq (30a)

y′(n+1)={hacek over (c)} ₀ ·{hacek over (x)}( n+1)+p ² ·y′(n−1),  Eq (30b)

where {hacek over (c)}₀=1| real (p²)|−|imag (p²)| and {hacek over (x)}(n) is the output of the FIR part.

The FIR part may be expressed as:

{hacek over (x)}(n)={hacek over (c)} ₀ ⁻¹ ·{tilde over (c)} ₀ ·x(n)+{hacek over (c)} ₀ ⁻¹ ·{tilde over (c)} ₀ ·p·x(n−1),and  Eq (31a)

{hacek over (x)}(n+1)={hacek over (c)} ₀ ⁻¹ ·{tilde over (c)} ₀ ·x(n+1)+{hacek over (c)} ₀ ⁻¹ ·{tilde over (c)} ₀ ·p·x(n).  Eq (31b)

FIG. 14 shows a block diagram of a complex first-order IIR section 1400 implemented with a FIR filter bank 1410 and an IIR filter bank 1420. FIR filter bank 1410 may filter the complex input samples x_(re)(n) and x_(im)(n) as shown in equation set (31) and provide complex filtered samples {hacek over (x)}_(re)(n) and {hacek over (x)}_(im)(n). IIR filter bank 1420 may filter the complex filtered samples {hacek over (x)}_(re)(n) and {hacek over (x)}_(im)(n) as shown in equation set (30) and provide complex output samples y_(re)′(n) and y_(im)′(n). FIG. 14 shows a design with IIR filter bank 1420 being placed after FIR filter bank 1410. In another design, FIR filter bank 1410 may be placed after IIR filter bank 1420. The order of FIR filter bank 1410 and IIR filter bank 1420 may be selected based on noise and other considerations.

FIG. 15 shows a schematic diagram of a design of a PSC filter 1500 that implements IIR filter bank 1420 in FIG. 14. PSC filter 1500 includes a first IIR section 1510 a and a second IIR section 1510 b. Each IIR section 1510 includes all elements of PSC filter 1000 in FIG. 10.

A real input signal V_(in,re) is provided to both a switch 1512 a in IIR section 1510 a and a switch 1512 c in IIR section 1510 b. An imaginary input signal V_(in,im) is provided to both a switch 1512 b in IIR section 1510 a and a switch 1512 d in IIR section 1510 b. A switch 1516 a in IIR section 1510 a and a switch 1516 c in IIR section 1510 b are coupled together and provide a real output signal V_(out,re). A switch 1516 b in IIR section 1510 a and a switch 1516 d in IIR section 1510 b are coupled together and provide an imaginary output signal V_(out,im). The other elements within IIR sections 1510 a and 1510 b are coupled as described above for FIG. 10.

IIR section 1510 a operates in each even-numbered clock cycle, filters the V_(in,re) and V_(in,im) signals, and provides the V_(out,re) and V_(out,im) signals. IIR section 1510 a is disabled during each odd-numbered clock cycle. Conversely, IIR section 1510 b operates in each odd-numbered clock cycle, filters the V_(in,re) and V_(in,im) signals, and provides the V_(out,re) and V_(out,im) signals. IIR section 1510 b is disabled during each even-numbered clock cycle. IIR sections 1510 a and 1510 b thus operate in an interleaved manner, with IIR section 1510 a operating in one clock cycle, then IIR section 1510 b operating in the next clock cycle, then IIR section 1510 a operating in the following clock cycle, etc. IIR section 1510 a operates on {hacek over (x)}(n) and provides y′(n), as shown in equation (30a). IIR section 1510 b operates on {hacek over (x)}(n+1) and provides y′(n+1), as shown in equation (30b).

In one design, a PSC filter for FIR filter bank 1410 in FIG. 14 (which may be a FIR filter) may be implemented in similar manner as PSC filter 1000 in FIG. 10. However, the switches for the PSC filter for FIR filter bank 1410 are operated to implement a FIR filter instead of an IIR filter. In another design, two FIR filter banks may be merged from two first-order complex IIR filters. The merged second-order FIR filter has real coefficients and may be implemented as a normal second-order PSC FIR filter.

Equation sets (30) and (31) show filter bank transformation for a case in which IIR filter bank 1420 includes two IIR sections. In general, the filter bank transformation may be performed for any value of m or M=2^(m) to obtain a complex pole at p² ^(m) =p^(M). The IIR part may be expressed as:

y′(n+i)={hacek over (c)}₀ ·{hacek over (x)}(n+i)+p ^(M) ·y′(n−M+i),for i=0, . . . ,M−1,  Eq (32)

where {hacek over (c)} ₀=1|real(p ^(M))|−imag(p ^(M))|.  Eq (33)

A filter bank may thus include M IIR sections. Each IIR section may be implemented as shown in FIG. 15, may operate at 1/M of the clock rate, and may be enabled every M clock cycles. The M IIR sections may be sequentially enabled in M clock cycles, one IIR section in each clock cycle.

FIG. 16 shows a plot 1610 of pole movement due to filter bank transformation with different values of M. An IIR filter is stable if its pole has a magnitude of r<1 and is located within a unit circle 1612. The pole of the IIR filter meets the power constraint in equation (20) if it is located within a diamond box 1614. The filter bank transformation changes the pole from p to p^(M). Plot 1610 shows the pole location for an example IIR filter with c₀=1, c₁=1.5 and c₂=0.6875. In this example, the pole p obtained from decomposition is located outside diamond box 1614 and thus does not meet the power constraint. The filter bank transformation with M=2 results in the pole p² being located within diamond box 1614 and thus meets the power constraint. The filter bank transformation with M=4 and 8 results in the poles p⁴ and p⁸ being located closer to the origin. As shown by this example, the pole generally moves toward the origin for larger values of M. In general, a larger distance from the pole to the unit circle may result in smaller insertion loss, which is desirable.

The pole due to filter bank transformation with M=2 may be expressed as:

p ² =r ² ·e ^(j2θ) ≡{hacek over (p)}={hacek over (p)} _(re) +j{hacek over (p)} _(im).  Eq (34)

The power constraint for the IIR part may then be expressed as:

|{hacek over (p)}| _(re) |+|{hacek over (p)} _(im) |=r ²·(|cos 2 θ|+|sin 2 θ|)≦1  Eq (35)

Since r<1 for a stable IR filter, r²<r and it may be more possible to meet the power constraint. As an example, a complex first-order IIR section with pole of p=0.625+j 0.625 does not meet the power constraint. However, a filter bank with pole of p²=j 0.78 meets the power constraint.

Equation (35) is for a 2-way interleaved filter bank with M=2. In general, the power constraint for an M-way interleaved filter bank may be expressed as:

|{hacek over (p)} _(re) |+|{hacek over (p)} |=r ^(M)·(|cos M·θ|+|sin M·θ|)≦1.  Eq (36)

In theory, the power constraint can always be satisfied if M is sufficiently large. However, a larger M also corresponds to greater complexity for the IIR part.

FIG. 17 shows a process 1700 for designing a PSC filter with filter bank transformation. A filter transfer function may be decomposed into a FIR part and an IIR part (block 1712). In one design, the filter transfer function is for a complex first-order IIR filter and may be decomposed into the FIR part and the IIR part. A complex coefficient (e.g., p^(M)) for the IIR part may be determined based on a complex coefficient (e.g., p) for the complex first-order IIR filter. The IIR part may be partitioned into multiple (M) IIR sections, with each IIR section operating at 1/M clock rate, and the M IIR sections being sequentially enabled in M clock cycles (block 1714). In one design, the IIR part may be partitioned into first and second IIR sections, with the first IIR section being enabled in even-numbered clock cycles, and the second IIR section being enabled in odd-numbered clock cycles. The FIR and IIR parts may be implemented with PSC filter sections to obtain the filter transfer function (block 1716).

In a fourth scheme for meeting the power constraint for IIR filter, pole repositioning may be performed, and a pole may be moved to a more suitable location in order to meet the power constraint. The second part of the right hand side of equation (22) may be expressed as:

f(θ)=(|cos θ|+|sin θ|).  Eq (37)

FIG. 18 shows a plot 1810 of function ƒ(θ) in equation (37). Function ƒ(θ) has a minimum value of 1.0 for θ=i·π/2, where i is an integer. In order to meet the power constraint |p_(re)|+|p_(im)|<1, it is better to minimize f(θ), which means making either θ or M B close to i·π/2. It may be desirable to have θ≈π/2, so that the poles are on the imaginary axis. If the interleaved filter bank with M=2 is used, then it may be desirable to have θ≈π/4 or θ≈3π/4. This may be achieved by a multi-rate filter design.

For the fourth scheme, the pole location may be varied in a systematic or pseudo-random manner. The pole at each new location may be evaluated to determine whether (i) a desired filter response can be obtained with the new pole location and (ii) the power constraint is met with the new pole location.

FIG. 19 shows a process 1900 for designing a second-order IIR filter to meet power constraint. The second-order IIR filter may be designed to meet applicable system requirements. The coefficients c₀, c₁ and c₂ of the second-order IIR filter may be obtained (block 1912). A determination is made whether the coefficients can be scaled, e.g., whether |c₁|+|c₂|<1 (block 1914). If the answer is ‘Yes’ for block 1914, then coefficient scaling may be performed to obtain coefficient c₀′ as shown in equation (15) (block 1924). Otherwise, the second-order IIR filter may be decomposed into complex first-order IIR sections (block 1916). A determination is made whether the pole of the first-order IIR sections meet the power constraint, e.g., whether |p_(re)|+|p_(im)|<1 (block 1918). If the answer is ‘Yes’ for block 1918, then coefficient scaling may be performed to obtain coefficient {tilde over (c)}₀ as shown in equation (22) (block 1924).

Otherwise, each complex first-order IIR section may be implemented with a filter bank starting with m=1 (block 1920). A determination is made whether the pole for the filter bank meets the power constraint, e.g., whether |{hacek over (p)}_(re)|+|{hacek over (p)}_(im)|<1 (block 1922). If the answer is ‘Yes’ for block 1922, then coefficient scaling may be performed to obtain coefficient {hacek over (c)}₀ as shown in equation (33) (block 1924). Otherwise, a determination is made whether m is equal to a maximum value (block 1926). If the answer is ‘No’, then m may be incremented (block 1928), and the process may then return to block 1920. Otherwise, pole repositioning may be performed in order to meet the power constraint (block 1930).

For clarity, much of the description above is for second-order FIR filter and first-order and second-order IIR filters. The PSC filters described herein may be used for FIR filters and IIR filters of any order.

The PSC filters described herein may provide certain advantages. First, the PSC filters do not utilize amplifiers within the PSC filters, which may reduce size and power consumption. Amplifiers may be used for input/output buffering. Second, the PSC filters may be able to provide an accurate frequency response, which is determined by capacitor ratios that can be more accurately achieved in an integrated circuit (IC). Third, the PSC filters may have high adaptability since it uses an array of capacitors that may be configured during operation, e.g., to obtain different filter responses.

The PSC filters described herein may be used for various applications such as wireless communication, computing, networking, consumer electronics, etc. The PSC filters may also be used for various devices such as wireless communication devices, cellular phones, broadcast receivers, personal digital assistants (PDAs), handheld devices, wireless modems, laptop computers, cordless phones, Bluetooth devices, wireless local loop (WLL) stations, consumer electronics devices, etc. For clarity, the use of the PSC filters in a wireless communication device, which may be a cellular phone or some other device, is described below. The PSC filters may be used to pass a desired signal, to attenuate jammers and out-of-band noise and interference, and/or to perform other functions in the wireless device.

FIG. 20 shows a block diagram of a design of a wireless communication device 2000 in which the PSC filter described herein may be implemented. Wireless device 2000 includes a receiver 2020 and a transmitter 2040 that support bi-directional communication. In general, wireless device 2000 may include any number of receivers and any number of transmitters for any number of communication systems and frequency bands.

In the receive path, an antenna 2012 may receive radio frequency (RF) modulated signals transmitted by base stations and provide a received RF signal, which may be routed through an RF unit 2014 and provided to receiver 2020. RF unit 2014 may include an RF switch and/or a duplexer that can multiplex RF signals for the transmit and receive paths. Within receiver 2020, a low noise transconductance amplifier (LNTA) 2022 may amplify the received RF signal (which may be a voltage signal) and provide an amplified RF signal (which may be a current signal). A passive sampler 2024 may sample the amplified RF signal, perform frequency downconversion via a sampling operation, and provide analog samples. An analog sample is an analog value for a discrete time instant. A filter/decimator 2026 may filter the analog samples, perform decimation, and provide filtered samples at a lower sample rate. Filter/decimator 2026 may be implemented with the PSC filters described herein.

The filtered samples from filter/decimator 2026 may be amplified by a variable gain amplifier (VGA) 2028, filtered by a filter 2030, further amplified by an amplifier (AMP) 2032, further filtered by a filter 2034, and digitized by an analog-to-digital converter (ADC) 2036 to obtain digital samples. Filter 2030 and/or 2034 may be implemented with the PSC filters described herein. VGA 2028 and/or amplifier 2032 may be implemented with switched-capacitor amplifiers that can amplify the analog samples from filters 2026 and 2030. A digital processor 2050 may process the digital samples to obtain decoded data and signaling. A control signal generator 2038 may generate a sampling clock for passive sampler 2024 and control signals for filters 2026, 2030 and 2034.

In the transmit path, transmitter 2040 may process output samples from digital processor 2050 and provide an output RF signal, which may be routed through RF unit 2014 and transmitted via antenna 2012. For simplicity, details of transmitter 2040 are not shown in FIG. 20.

Digital processor 2050 may include various processing units for data transmission and reception as well as other functions. For example, digital processor 2050 may include a digital signal processor (DSP), a reduced instruction set computer (RISC) processor, a central processing unit (CPU), etc. A controller/processor 2060 may control the operation at wireless device 2000. A memory 2062 may store program codes and data for wireless device 2000. Data processor 2050, controller/processor 2060, and/or memory 2062 may be implemented on one or more application specific integrated circuits (ASICs) and/or other ICs.

FIG. 20 shows a specific design of receiver 2020. In general, the conditioning of the signals within receiver 2020 may be performed by one or more stages of mixer, amplifier, filter, etc. These circuit blocks may be arranged differently from the configuration shown in FIG. 20. Furthermore, other circuit blocks not shown in FIG. 20 may also be used to condition the signals in the receiver. Some circuit blocks in FIG. 20 may also be omitted. All or a portion of receiver 2020 may be implemented on one or more RF ICs (RFICs), mixed-signal ICs, etc.

The received RF signal from antenna 2012 may contain both a desired signal and jammers. A jammer is a large amplitude undesired signal close in frequency to a desired signal. The jammers may be attenuated prior to ADC 2036 in order to avoid saturation of the ADC. Filters 2026, 2030 and/or 2034 may attenuate the jammers and other out-of-band noise and interference and may each be implemented with any of the PSC filters described herein.

The PSC filters described herein may be implemented on an IC, an analog IC, an RFIC, a mixed-signal IC, an ASIC, a printed circuit board (PCB), an electronics device, etc. The PSC filters may also be fabricated with various IC process technologies such as complementary metal oxide semiconductor (CMOS), N-channel MOS (NMOS), P-channel MOS (PMOS), bipolar junction transistor (BJT), bipolar-CMOS (BiCMOS), silicon germanium (SiGe), gallium arsenide (GaAs), etc.

An apparatus implementing the PSC filters described herein may be a stand-alone device or may be part of a larger device. A device may be (i) a stand-alone IC, (ii) a set of one or more ICs that may include memory ICs for storing data and/or instructions, (iii) an RFIC such as an RF receiver (RFR) or an RF transmitter/receiver (RTR), (iv) an ASIC such as a mobile station modem (MSM), (v) a module that may be embedded within other devices, (vi) a receiver, cellular phone, wireless device, handset, or mobile unit, (vii) etc.

In one or more exemplary designs, the functions described may be implemented in hardware, software, firmware, or any combination thereof. If implemented in software, the functions may be stored on or transmitted over as one or more instructions or code on a computer-readable medium. Computer-readable media includes both computer storage media and communication media including any medium that facilitates transfer of a computer program from one place to another. A storage media may be any available media that can be accessed by a computer. By way of example, and not limitation, such computer-readable media can comprise RAM, ROM, EEPROM, CD-ROM or other optical disk storage, magnetic disk storage or other magnetic storage devices, or any other medium that can be used to carry or store desired program code in the form of instructions or data structures and that can be accessed by a computer. Also, any connection is properly termed a computer-readable medium. For example, if the software is transmitted from a website, server, or other remote source using a coaxial cable, fiber optic cable, twisted pair, digital subscriber line (DSL), or wireless technologies such as infrared, radio, and microwave, then the coaxial cable, fiber optic cable, twisted pair, DSL, or wireless technologies such as infrared, radio, and microwave are included in the definition of medium. Disk and disc, as used herein, includes compact disc (CD), laser disc, optical disc, digital versatile disc (DVD), floppy disk and blu-ray disc where disks usually reproduce data magnetically, while discs reproduce data optically with lasers. Combinations of the above should also be included within the scope of computer-readable media.

The previous description of the disclosure is provided to enable any person skilled in the art to make or use the disclosure. Various modifications to the disclosure will be readily apparent to those skilled in the art, and the generic principles defined herein may be applied to other variations without departing from the scope of the disclosure. Thus, the disclosure is not intended to be limited to the examples and designs described herein but is to be accorded the widest scope consistent with the principles and novel features disclosed herein. 

What is claimed is:
 1. A method comprising: obtaining multiple coefficients for a filter transfer function; scaling at least one of the multiple coefficients based on a power constraint for a passive switched-capacitor (PSC) filter; and implementing the PSC filter based on the at least one scaled coefficient to obtain the filter transfer function.
 2. The method of claim 1, wherein the filter transfer function is for a finite impulse response (FIR) filter, and wherein the scaling at least one of the multiple coefficients comprises determining a scaling factor based on magnitude of each of the multiple coefficients, and scaling each of the multiple coefficients based on the scaling factor to obtain a corresponding scaled coefficient.
 3. The method of claim 2, wherein the filter transfer function is for a second-order FIR filter, and wherein the power constraint comprises |b ₀ ′|+|b ₁ ′|+|b ₂′|=1, where b₀″, b₁′ and b₂′ are three scaled coefficients for the second-order FIR filter.
 4. The method of claim 1, wherein the filter transfer function is for an infinite impulse response (IIR) filter, and wherein the scaling at least one of the multiple coefficients comprises replacing one of the multiple coefficients with a new coefficient determined based on magnitude of each remaining coefficient.
 5. The method of claim 4, wherein the filter transfer function is for a second-order IIR filter, and wherein the power constraint comprises |c ₀ ′|+|c ₁ |+|c ₂|=1, where c₀′ is the new coefficient and c₁ and c₂ are two coefficients for the second-order IIR filter.
 6. An apparatus comprising: means for obtaining multiple coefficients for a filter transfer function; means for scaling at least one of the multiple coefficients based on a power constraint for a passive switched-capacitor (PSC) filter; and means for implementing the PSC filter based on the at least one scaled coefficient to obtain the filter transfer function.
 7. The apparatus of claim 6, wherein the filter transfer function is for a finite impulse response (FIR) filter, and wherein the means for scaling at least one of the multiple coefficients comprises means for determining a scaling factor based on magnitude of each of the multiple coefficients, and means for scaling each of the multiple coefficients based on the scaling factor to obtain a corresponding scaled coefficient.
 8. The apparatus of claim 6, wherein the filter transfer function is for an infinite impulse response (IIR) filter, and wherein the means for scaling at least one of the multiple coefficients comprises means for replacing one of the multiple coefficients with a new coefficient determined based on magnitude of each remaining coefficient.
 9. A computer program product, comprising: a computer-readable medium comprising: code for causing at least one computer to obtain multiple coefficients for a filter transfer function; code for causing the at least one computer to scale at least one of the multiple coefficients based on a power constraint for a passive switched-capacitor (PSC) filter; and code for causing the at least one computer to implement the PSC filter based on the at least one scaled coefficient to obtain the filter transfer function.
 10. A method comprising: decomposing a filter transfer function into multiple complex first-order filter sections; and implementing the multiple complex first-order filter sections with multiple passive switched-capacitor (PSC) filter sections to obtain the filter transfer function.
 11. The method of claim 10, wherein the decomposing the filter transfer function comprises decomposing the filter transfer function for a second-order infinite impulse response (IIR) filter into two complex first-order IIR sections.
 12. The method of claim 11, wherein the decomposing the filter transfer function further comprises determining complex coefficients for the two complex first-order IIR sections based on coefficients for the filter transfer function.
 13. The method of claim 12, wherein the decomposing the filter transfer function further comprises determining an input coefficient for the two complex first-order IIR sections based on magnitude of real and imaginary parts of the complex coefficients.
 14. A method comprising: decomposing a filter transfer function into a finite impulse response (FIR) part and an infinite impulse response (IIR) part; and implementing the FIR part and the IIR part with passive switched-capacitor (PSC) filter sections to obtain the filter transfer function.
 15. The method of claim 14, wherein the decomposing the filter transfer function comprises decomposing the filter transfer function for a complex first-order IIR filter into the FIR part and the IIR part, and determining a complex coefficient for the IIR part based on a complex coefficient for the complex first-order IIR filter.
 16. The method of claim 15, wherein the complex coefficient for the IIR part is p^(M), where p is the complex coefficient for the complex first-order IIR filter and M is an integer greater than one.
 17. The method of claim 14, wherein the decomposing the filter transfer function comprises partitioning the IIR part into multiple (M) IIR sections, each IIR section operating at 1/M clock rate, the M IIR sections being sequentially enabled in M clock cycles.
 18. The method of claim 14, wherein the decomposing the filter transfer function comprises partitioning the IIR part into first and second IIR sections, the first IIR section being enabled in even-numbered clock cycles, and the second IIR section being enabled in odd-numbered clock cycles. 